Low Loss Tunable Filters in Substrate Integrated Waveguide

Abstract

In this work, the design and experimental results of com-

pact low-loss filters, based on the extension of the clas-

sical coaxial waveguide resonator to Substrate Integrated

Waveguide technology, are successfully demonstrated.

The design, fabrication and measurement of a conti-

nuously tunable cavity resonator and a two-pole filter at

S-band in single-layer SIW technology are presented.

These structures keep the low-cost fabrication scheme of

single-layer PCB processing, while requiring less than half

the area compared to a conventional SIW design.

Keywords:Tunable filters, varactor tuning diodes, com-

bline resonator, compact filter design, microwave filter,

continuous tuning, Substrate Integrated Waveguide

(SIW). 1. Introduction The concept of Substrate Integrated Waveguide (SIW)has been developed to synthesize waveguide cavities with planar structures into a single dielectric material,combining integrated high Q-factor filters with low cost fabrication techniques. A SIW component consists on a synthetic waveguide formed by the top and the bottom metal plates of a dielectric slab and is bounded transver-sely by two sidewalls of cylindrical metallic via-holes. The rows of vias connect top and bottom metal layers for-ming the cavity walls. Input and output coupling is achie-ved using tapered microstrip transmission lines or coplanar current probes. Adjusting the geometrical di-mensions of the taper or probes, the input/output cou-pling can be controlled.

The SIW components can be fabricated on printed circuit boards (PCBs) or low-temperature co-fired ceramics (LTCCs) technologies. They enable a significant reduction of size, weight and cost compared with metallic wave-guide designs. As the SIWs are based in the TE n0modes,the characteristic of conventional waveguides are preser-ved and the propagation of energy of these modes is subs-tantially limited in the substrate. Thus, they have a higher Q-factor and lower loss than other planar guided-wave structures, for instance microstrip lines and coplanar wa-veguides (CPW). In fact, conventional planar resonators provide a moderate unloaded Q-factor, typically less than Q u <100-200, while SIW resonators can reach a Q-factor higher than 500 using low-loss substrates. Furthermore, a growing interest has also focused on SIW technology be-cause they can be easily integrated with microstrip or co-planar waveguide circuits [1] on the same laminate.

Traditionally, the adaptive function of the front-end prese-

lection filters is achieved using a bank of switchable fixed

frequency filters for multiple frequency bands [2]. Howe-

ver, this means that both the size and cost increase, due

to the complexity of the filter structures. Recently, due to

their potential to reduce system size and complexity, tuna-

ble filters are called to be key components in the future

microwave communication subsystems, particularly in the

development of future multi-band multi-standard minia-

turized front-ends. This trend has led to the increased de-

mand for filter technologies with frequency tunable ability.

Therefore, in the design of low-loss reconfigurable filters,

the availability of high-Q tunable resonators is of primary

concern. While frequency and bandwidth tunability have

been extensively demonstrated in planar technology using

different tuning elements (e.g. varactor diodes [3], ferroe-lectrics [4] or MEMS [5, 6]), an increasing interest has 69Waves - 2012 - year 4/ISSN 1889-8297Low Loss Tunable Filters in Substrate Integrated Waveguide

S. Sirci, J. D. Martínez, M. Taroncher and Vicente E. Boria

Instituto de Telecomunicaciones y Aplicaciones Multimedia,

Universitat Politècnica de València,

8G Building - access D - Camino de Vera s/n - 46022 Valencia (Spain)

Corresponding author: ssirci@iteam.upv.es

grown on electronically tunable SIW cavity resonators as an enabling technology for the design of very low-loss tu-nable filters [7, 8] that can be an alternative to filter banks in several applications.

In this paper we present a novel structure for implemen-

ting high-Q frequency agile resonators based on com-bline SIW cavities loaded with GaA s varactor tuning diodes. Following this concept, the design of compact and continuously tunable filters based on a combline to-pology is presented. The resonators and filters are fabri-cated in a low-cost microwave substrate using a conventional single-side PCB fabrication process and off-the-shelf GaAs varactors.

Some previous research has been very recently done in the study of reconfigurable SIW structures [9-11], con-cerning mainly the design of discretely tunable filters. A first attempt of using perturbing conductive elements for tuning the resonant frequency of a SIW cavity was pro-posed in [9], although no experimental device was then reported. More recently, a PIN diode-based discretely tu-nable SIW resonator consisting on switchable perturbing elements has been proposed [10]. Finally, a different ap-proach for center frequency tuning of multi-layer filters has made use of relatively large piezoelectric actuators disks with remarkable results, however at the price of a more sophisticated design [11].

In this paper we propose a continuously tunable SIW filter based on a combline topology. Then, an S-band combline filter is successfully designed, fabricated and measured. Ex-perimental and simulation results are in good agreement, showing the potential advantages of this structure in terms of size and design flexibility, without increasing insertion losses compared to its conventional SIW counterpart. The main advantages of our structure are the following: con-tinuous tuning range, more compact design, negligible power consumption and low-cost fabrication process. Mo-reover, a model for the proposed resonator is considered, and a systematic procedure for the design of these minia-turized coupled resonator bandpass filters is presented.

2. Combline Tunable Resonator in SIW

Technology

2.1 Resonator Structure

The structure of the tunable combline SIW resonator is shown in Fig. 1. It consists on a square cavity (size a) where a conducting post has been inserted at the center of the resonator using a plated via hole. A metal patch connected to that post and separated from the SIW top plate by a small gap is created on the top side of the ca-vity. So, the inner via is short-circuited at the bottom me-tallic plan and open ended at the top. A loading capacitance C0is therefore established between the post and the top metal layer of the SIW through the fringing

fields across the annular gap.

Thus, the proposed resonator can be seen as a square-circular coaxial waveguide cavity, and can be modeled as a piece of TEM mode transmission line short-circuited at one end and capacitively terminated at the other. A scheme of the equivalent circuit is shown in Fig. 2.

Due to the previous model, the resonant frequency of the SIW combline resonator can be expressed as a function of the substrate thickness l, the annular gap capacitance C0and the characteristic impedance of the rectangular-circular coaxial waveguide Z0

.

In order to validate the proposed concept, we have stu-

died the tuning range and unloaded quality factor of a

tunable combline SIW resonator. Firstly, a device about 4

GHz has been designed using full-wave 3D simulation of

the structure using HFSS. The dimensions of the designed

resonator can be seen in Fig. 3. It is worth pointing out

that the excitation of the resonator is performed through

magnetic coupling through CPW probes.

As shown, the resonant frequency f R is not only a func-

tion of the width and length of the cavity, but also a func-70ISSN 1889-8297/Waves - 2012 - year 4

Figure 1.Drawing of the resonator, showing the con-

ducting post and the floating and metal patches. The availability of high-Q tunable resonators is of pri-

mary concern in the design of low-loss reconfigurable fil-

ters which are called to be key components in the future

microwave communication subsystems.

tion of size a of the metallic patch, the isolating gap and the diameter of the central conducting post. These pro-perties of the combline resonator can be used to reduce the resonant frequency, without a significant decreasing in the Q factor of the resonator.

Fig. 4(a) and 4(b) show the dependence of f

R and unloa-ded Q0factor on the dimensions of the central patch for the proposed combline resonator on ?r= 3.55 substrate. It is seen that a high percentage change of the f R can be obtained changing the length

a between 2 and 3 mm, with low degradation of Q0factor.

2.2 Tuning Element

The resonant frequency of the filter can be controlled by changing the loading capacitance C0of the combline re-sonator as shown in Fig. 5. Thus, GaAs varactor diodes have been inserted connected to the metal patch for mo-dulating the capacitance of the resonator.

A hyper-abrupt junction GaAs varactor has been used in this work as tuning element. Particularly, we have chosen

the MA46H200 from MA-COM. It exhibits a capacitance

variation from 0.25 to 1 pF for a reverse bias voltage bet-

ween 22 and 0 V respectively. The nominal Q?factor at

50 MHz is 3000 giving an estimated series resistance of

2.2 Ω

.

An important advantage of using tuning varactors ins-

tead of switching elements (i.e. PIN diodes) is the negli-

gible power consumption of the device while operating

in reverse bias conditions. Moreover, the need of only one

varactor per cavity significantly reduces the complexity of

the bias network.

2.3 Continuously Tunable Resonator

The loading capacitance C0, that would initially include

the capacitance between the metal patch and the SIW

top plate, can be controlled by inserting a varactor con-

necting both sides. In order to simplify the biasing and

optimizing the varactor RF performance, a floating metal

island (it can be seen in Fig. 5) is created between the

metal patch and the SIW top plate.

The cathode of the varactor is then connected to this flo-

ating pad while a series capacitor is inserted between the

former and the top metal side of the cavity for providing

a low impedance path for the microwave signal. The se-

ries DC blocking capacitor has been chosen of 1.5 pF in

order to slightly reduce the capacitive effect of the varac-

71 Waves - 2012 - year 4/ISSN 1889-8297

Figure 5.Drawing of the cross-section of the combline

resonator. The varactor is connected between the island

and the metal patch.

Figure 263e828b856a561252d36f7eyout of the designed resonator. Via hole

and conductive post diameters are 0.5 and 0.3 mm res-

pectively. Metal patch gap is 0.1 mm.

Figure 4.Change in resonant frequency fR (left) and

Q0 (right) as a function of the size a of metallic patch.

nator. We have chosen to keep the varactor bias circuit

as simple as possible, so a thin wire has been connected

from the floating metal island to a bias pad. Thus, a con-

venient bias voltage can be applied between the floating

pad and the top metal of the SIW cavity by using this thin

wire. DC bias and ground connections have been isolated

by using high value resistors (i.e. R>100 k Ω), specifically

1 M Ω resistors have been used.

Then, internal ports have been created at the places

where the varactor and the fixed capacitance will be in-

serted in order to combine 3D EM simulations with cir-

cuital simulations using the equivalent models of the

lumped components. The equivalent circuit model of the

tunable combline resonator is shown in Fig. 6, including

parasitic components for the varactor diode and the DC

blocking capacitance.

The resonator has been strongly under-coupled (i.e. the S 21parameter has been kept below -20 dB) in order to enable

us the accurate measurement of the unloaded Q factor

from the transmission response of the device [12]. Simula-

tions for different values of varactor capacitance have been The estimated unloaded Q 0ranges from 180 to 70 for a capacitance variation between 0.25 pF and 1.25 pF . For the same capacitance range, the resonance frequency

changes from 3.3 GHz to 2.8 GHz. Thus, an estimated tuning range about 15% can be extracted from the ob-tained results. The nominal parasitic due to the varactor packaging and the series capacitor have been included in the simulations. 3. Filter Design in SIW Technology 3.1 Combline Filter For high-Q filter, it is critical to achieve center frequency tuning without degrading the Q factor of the filter. Since

tunable lumped components are generally low Q at mi-crowave frequency, they will decrease the overall quality factor. The proposed tunable structure allows the possi-bility of employing low-Q varactors or high-Q MEMS va-ractors to achieve center frequency control, with slightly degradation of the overall Q , as shown in the previous chapter.

The structure of a 2?pole combline filter in SIW techno-

logy can be seen in Fig. 8. The filter consists of two tu-

nable SIW resonators, coupled by irises. In fact, narrow

n-order bandpass filters can be implemented by inserting

inductive windows coupling adjacent resonators.

Each resonator consists on a square/circular SIW cavity of

side a where a plated via hole has been inserted at the

center of the resonator, where the electromagnetic field

of the fundamental mode is maximum. This inner via is

short-circuited at the bottom side and connected to a

metal patch electrically isolated from the top side through

a small gap.

A capacitive effect is therefore established between this

metal patch and the top plane of the SIW. This capaci-

tance value depends on the perimeter of the metal patch,

the spacing of the gap and the thickness of the metalli-

zation layer.72ISSN 1889-8297/

Waves - 2012 - year 4Figure 6.The equivalent circuit model of the tunable

resonator used in circuit simulations.

Figure 7.Simulated results of the under-coupled tunable

SIW resonator for different values of varactor capacitance.

For the input/output couplings to the resonators, we have maintained the same structure of previous combline resonator. These are performed by means of coplanar waveguide CPW probes as shown in Fig.8. Therefore, the external coupling can be therefore controlled by the ge-ometrical dimensions of the probes. The inter-resonators coupling is controlled by means of irises created on the adjacent walls. Since the diameter of the rows of via holes is considered fixed, the coupling level can be ad-justed changing the size of the iris and the distance bet-ween iris via holes.

As shown, the proposed tunable SIW resonator can be seen as a TEM square-circular coaxial transmission line of length h (i.e. the substrate height) short-circuited at one end and capacitively terminated at the other. The suscep-tance ?(?) of the coaxial resonator can be expressed as [1]

Where, C 0is the total capacitance of the central metal patch, Y 0the characteristic admittance of the TEM mode short-circuited transmission line formed by the inner (cir-cular) and outer (rectangular) conductor, h the substrate thickness and the phase constant of the coaxial line at frequency ?is

[2]Where, ?r is the dielectric permittivity of the substrate,while C 0is the speed of light in vacuum. Now, the reso-nant frequency of the SIW cavity can be computed from the equation ?(?)=0, that is [3]

The inductive contribution comes from the TEM mode short-circuited resonator, which can be seen as a short piece of coaxial transmission line of length and charac-teristic admittance embedded into the dielectric material of permittivity ?r . For a circular inner conductor of dia-meter d and a square contour of side a , the characteristic admittance can be well approximated when a>>d by [13][4]Finally, from [14], the capacitance established between the metal disk of radius and the SIW top metal can be computed. It is worth mentioning, the thickness of the

metallic layers is not considered in this expression.

3.2 Filter Design

The synthesis and dimensioning of the tunable filter is per-formed by the classical approach for directly coupled reso-nator filters. Firstly, the low-pass prototype parameters g 0,g 1,…,g n+1are computed from the desired response. From these values, the external quality factor Q ext and inter-re-sonator coupling coefficients K ij are obtained from the well-known expressions [12]. Secondly, a SIW combline re-sonator at the center frequency of the filter is designed ta-king into account the dielectric constant ?r and the substrate thickness h . The side of the square SIW cavity and the diameter of the inner via will determine the Y 0value of the coaxial resonator. Additionally, the dimension of the metal patch and the isolating gap width control the loading capacitance C 0.

The prototype of the combline bandpass filter can be mo-deled using shunt resonators and frequency-invariant ad-mittance inverters, as shown in Fig. 9. Given a filter response with centre frequency and bandwidth, being and the upper and lower cutoff frequencies, the loading capacitance and the admittance of the coaxial resonator can be obtained by mapping the resonator susceptance and the admittance inverters of the bandpass filter with those of the low-pass prototype. The former condition can be expressed as [5]

Where b is the resonator slope parameter, ??and ?c = 1 rad/s the angular frequency and cut-off frequency of the low-pass prototype respectively. The level of b will impact the unloaded Q factor of the resonator, and must be a trade-off between low losses, compactness and fe-asibility of the synthesized values of C 0and Y 0.

Now, substituting (1) into (4) for the corresponding pass-band edge frequencies ??=± 1?={??, ?L }we can obtain [6]And 73Waves - 2012 - year 4/ISSN 1889-8297Figure 9.Equivalent prototype of an N th -order combline resonator bandpass filter with frequency invariant admit-tance inverters.

?(?)=?C 0-Y 0 cot(?h )?=???r C 0?C 0=Y 0 cot h ???r C 0Y 0=In a d ??

r 60 1.079-1?(?)

=??b ???c ?0C 0=

b ???0cot ??+cot ?L ??cot ?L -?L cot ??The proposed resonator can be seen as a square-circular coaxial waveguide cavity, and can be modeled as a piece of TEM mode transmission line short-circuited at one end and capacitively terminated at the other.

[7]

Finally, the admittance inverters can be computed from the low-pass prototype coefficients using the well-known expressions

[8]

For i = 1 to (n – 1), where Y A is the admittance of the input/output access ports.

Following this approach, a 2-pole Chebyshev filter cen-tered at 3.1 GHz with 5% fractional bandwidth will be synthesized. Once the SIW cavity resonator has been de-signed, the values of Q ext and K ij are computed as a func-tion of the input/output CPW probe dimensions and the width of the coupling irises using 3D full-wave simula-tions with HFSS software, as shown in Fig. 10. In order to match the desired response of the filter, the resonators are tuned from the reflection group delay res-

ponse of short-circuited cavities. For adjusting the centre frequency, the dimensions of the central square post or the size of the cavities are modified. Moreover, by chan-ging the diameter of the inner vias the central frequency can be coarsely adjusted.

A final optimization of the filter dimensions is then per-formed for the minimum capacitance value of the tuning element, combining 3D full-wave results with circuital si-mulations using the equivalent models of the lumped components. As previous exposed, in the 3D model of the circuit, internal ports have been created at the places where the varactors 46H200 (from MA-COM) and the fixed capacitance will be inserted.

Since all the parameters are now determined, the simu-lated response of the designed filter can be obtained using ANSOFT Designer and is shown is shown in Fig. 11.The minimum capacitance of the varactor (0.25 pF) has been chosen to run this simulation. The centre frequency of the complete bandpass filter is 2.88 GHz and the in-sertion loss is estimated less than 1.5 dB.

4 Experimental Results

4.1 Fabrication and Measurement of Tunable SIW Resonator

A prototype of the resonator has been fabricated using Ro-gers RT/Duroid 4003C laminate (?r = 3.55 and tan ?= 2.7 ·10?3). A picture of the fabricated device including the biasing wires is shown in Fig. 12. The size of the reso-nator is 27 × 27 mm 2, showing an important size reduction compared to its equivalent conventional SIW counterpart due to the capacitive loading of the metal patch.The measured S 21parameters for the tunable combline SIW resonator are shown in Fig. 13. A tuning range of almost 20% is observed when changing the reverse bias voltage of the varactor from 2 to 22 V .

74ISSN 1889-8297/Waves - 2012 - year 4

Y 0=b

???0

??+?L

??cot ?L -?L cot ??

J 0,1=

?

b

???0

Y A g 0g 1J n,n+1=

?

b

???0

Y A g n g n+1

J i,i+1=?1g i g i+1

b

??

?0

Figure 11.Simulated results of the design filter (thick)and ideal response (dashed) for a minimum varactor ca-pacitance of 0.25 pF

.

The resonant frequency and the unloaded Q of the reso-nator as a function of the reverse bias voltage are shown in Fig. 14. As can be seen, the tunable Q of the resonator ranges from 40 to 150, being above 100 for the first 200 MHz of frequency shift. The power consumption of the device is negligible along the whole tuning voltages (i.e. typically less than 1 ?W). Although, the percentage value of tuning range is similar to estimated value, the range of frequency has moved to lower frequencies, since it ranges from 3.1 GHz to 2.6 GHz. The differences with the simulated results are attributed to an underestimation of the capacitance bet-ween the metal patch and the floating island where the

DC bias is applied. This effect slightly increases the total loading capacitance, decreasing both the resonant fre-quency of the structure and the achievable tuning range of the resonator. Fabrication and permittivity tolerances contribute also to the small shift to lower frequencies of the response. The small increase of losses between expe-rimental and simulated results is attributed to the moun-ting and soldering of the components. 4.2 Fabrication and Measurement of Tunable SIW 2-

pole Filter

To validate our filter topology, a 2?pole bandpass Chebyshev filter was designed, fabricated and measured in 1.524 mm-thick Rogers RO4003C substrate. The fixed combline filter is first designed with equiripple fractional bandwidth of 5%, center frequency of 3.1 GHz while the requested in-band return losses are ?20 dB. Simulation results of the tunable filter for the whole varactor capa-citance range can be seen in Fig. 15.

The prototype was fabricated using standard single-layer PCB processing technology. The layout and dimensions of the fabricated filter is depicted in Fig. 16. Filter size are 55×27.5 mm 2, while the equivalent conventional SIW fil-ter would require about twice the area for square cavities using the dominant TE101 mode.75Waves - 2012 - year 4

/ISSN 1889-8297

Figure 14.Resonance frequency (right) and unloaded Q (left) of the tunable resonator as a function of the re-

verse voltage applied to the varactor.

Figure 13.Simulated results of the under-coupled tunable SIW resonator for different values of varactor capacitance.

The input/output CPW lines present a trace width of

1.3 mm and a ground plane spacing of 0.15 mm, so the

characteristic impedance is closed to 50 Ω. The diameter of the vias for the SIW cavities and irises is 600 ?m , and the spacing between the contour vias is 1.25 mm. The central via hole presents a diameter of 300 ?m and the metal patch isolating gap is 100 ?m . Photography of the fabricated filter and the measured S-parameters are shown in Fig. 17.Measured results agree quite well with the simulations.The device centre frequency can be tuned between 2.64and 2.88 GHz for a bias voltage ranging from 3.5 – 22 V .The insertion losses vary from 1.27 dB to 3.63 dB across the whole tuning range. Moreover, a variation of the

3?dB fractional bandwidth between 4 %?7 % has been observed due to the short length of the coaxial resonator.

The tuning range of the proposed approach is mainly li-mited by the moderate

Q

-factor of the tuning elements

and the frequency dependence of the input/output and

inter-resonators couplings. However, the more compact

design would enable us to integrate several switchable reconfigurable filters in order to cover a broader fre-

quency band. Moreover, the use of higher Q

tuning ele-

ments (e.g. MEMS switches and varactors) will allow an

increase of the tuning range without degrading the EM performance of the device.The presence of the varactor introduces non-linearity ef-

fects that limit the power handling of the filter. Results

of intermodulation distortion and linearity of the fabrica-ted filter can be seen in Fig. 18. It is worth mentioning 76ISSN 1889-8297/Waves - 2012 - year 4Main advantages of combline SIW tunable filter are the

following: continuous tuning range, negligible power

consumption, more compact design and low-cost fabri-

cation process.

Figure 17.(Left): Photography of the tunable filter during the reconfiguration time measurement. (Right): Measured

S-parameters of the fabricated tunable 2-pole filter.Figure 16.(Left): Photography of the tunable filter during the 1-dB compression point measurement. (Right): Layout and dimensions of the fabricated tunable filter.

that these effects have been measured for the worst-case

of a high varactor capacitance.

Thus, a two-tone third-order intermodulation intercept

point (IIP3) of +24 dBm has been measured for a bias vol-

tage V b = 5 V and a two-tone separation ?f = 50 kHz.

However, the 1?dB compression point of the filter at the

same bias voltage is only +10 dBm.

The reconfiguration time of the filter has been measured

using a crystal power detector at the filter output and a

4 dBm tone centered at 2.8

5 GHz at the input. Then, the

bias voltage has been changed between 20 and 14 V

with a 100?s -period square signal. The measurement

setup is shown in Fig. 17. The low-to-high and high-to-

low measured reconfiguration times of the filter are 120

and 720 ns for a centre frequency shift about 40 MHz.5 Conclusions

A tunable combline SIW resonator has been proposed in

this paper. The capacitively loaded end of the resonator

has been used in order to include a surface-mounted tu-

ning varactor diode that changes the resonant frequency

of the device. A Q -factor better than 100 has been ob-

tained for a 200 MHz tuning range from 3.1 GHz. The

proposed device can be fabricated using a low-cost PCB

process and using off-the-shelf GaAs varactor diodes. The

structure presents a continuous tuning range of almost

20% and negligible power consumption.

The obtained results are very promising compared to typi-

cal planar tunable resonators [3, 5, 6], while other SIW

tunable cavities based on semiconductor tuning elements

have achieved unloaded Q ’s around 100 on a discretely

tunable approach [10]. Moreover, losses due to the tu-

ning element could be significantly reduced by using RF

MEMS varactors.

A continuously tunable filter has been designed, fabrica-

ted and measured in SIW technology. Main advantages

of the proposed structures are analog tuning range, low

losses and easy integration with other planar circuits. Ad-

ditionally, due to the combline topology of the devices, a very compact implementation is also achieved with a very wide spurious-free band. This approach shows inte-resting applications for the design of low-loss tunable fil-ters based on compact high-Q SIW resonators. Moreover,the application of this technology to the post-manufac-turing tuning of narrow-band SIW filters could be studied showing promising results. The design, performance and manufacturability of novel combline filters in SIW tech-nology have been demonstrated. The measured results show an excellent agreement with the EM simulations when all involved physical effects are considered. The proposed topology presents important advantages in terms of size compactness, spurious rejection and design flexibility.Acknowledgments

The authors would like to thank Generalitat Valenciana and

MICINN (Spanish Government) for its financial support

under projects GV/2009/007 and TEC2010-21520-C04-01.References [1] D. Deslandes and K. Wu, “Single-substrate integra-tion technique of planar circuits and waveguide fil-ters”, IEEE Trans. Microwave Theory & Tech., vol. 51,no. 2, pp. 593-596, Feb. 2003.[2] D. Packiaraj, V.S. Reddy, G.J. Mello, and A.T. Kalg-hatgi, “Electronically switchable suspended substrate stripline filters”, in Proc. RF and Microwave Confe-rence, pp. 64-66, Oct 2004. [3] I.C. Hunter, and J.D. Rhodes, “Electronically tunable mi-crowave bandpass filters”, IEEE Trans. Microwave The-ory & T ech., vol. 30, no. 9, pp. 1354-1360, Sept. 1982.[4] J. Nath, D. Ghosh, J.P . Maria, A.I. Kingon, W. Fathel-bab, P .D. Franzon, and M.B. Steer, ”An electronically tunable microstrip Bandpass filter using thin-film ba-rium-strontium-titanate (BST) varactors”, IEEE Trans.Microwave Theory Tech., vol. 53, no. 9, pp. 2707-2712, Sept. 2005.[5] A. Pothier, J.-C. Orlianges, E. Zheng, C. Champeaux, A.Catherinot,D. Cros, P . Blondy, and J. Papapolymerou,”Low loss two bit tunable bandpass filters using MEMS 77Waves - 2012 - year 4

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Biographies

Stefano Sirci was born in Torino,Italy. He received the B.S. and M.S degrees in Electronic Engineering from the University of Perugia, Peru-gia, Italy, in 2006 and 2009, respec-tively. He is current working toward the PhD degree at the Universitat Po-litècnica de València, Valencia, Spain.

Since 2009, he has been with the Microwave Application Group (GAM) at the Institute of Telecommunications and Multimedia Applications (iTEAM) at the Universitat Poli-tècnica de València. His research interest is currently fo-cused on designing, fabrication and measurement of tunable microwave filter in substrate integrated wave-guide (SIW) technology.

Jorge D. Martínez received the In-geniero de Telecomunicación and the Doctor Ingeniero de Telecomu-nicación degrees from the Universi-tat Politècnica de València, Valencia,Spain, in 2002 and 2008. He joined the Department of Electronic Engi-neering of the Universitat Politècnica de València in 2002, where he is As-sociate Professor since 2009. He was a Visiting Researcher at XLIM (Centre National de la Recherche Scientifique /Université de Limoges) from June 2006 to September 2007, working on the design and fabrication of radio-fre-quency microelectromechanical systems (RF MEMS). His current research interests are focused on emerging tech-nologies for reconfigurable microwave components with emphasis on tunable filters and RF MEMS.

Máriam T aroncher (S’03) was born in Lliria, Valencia, Spain, on October 8, 1979. She received the T elecom-munications Engineering degree from the Universidad Politécnica de Valencia (UPV), Valencia, Spain, in 2003, and is currently working to-ward the Ph.D. degree at UPV .

From 2002 to 2004, she was a Fellow Researcher with the UPV . Since 2004, she has been a Technical Researcher in charge of the experimental laboratory for high power effects in microwave devices at the Research Institute iTEAM, UPV . In 2006 she was awarded a Trainee position at the European Space Research and Technology Centre,European Space Agency (ESTEC-ESA), Noordwijk, The Netherlands, where she worked in the Payload Systems Division Laboratory in the area of Multipactor, Corona Discharge and Passive Intermodulation (PIM) effects. Her current research interests include numerical methods for the analysis of waveguide structures and the acceleration of the electromagnetic analysis methods.

Vicente E. Boria received the Inge-niero de Telecomunicación and the Doctor Ingeniero de Telecomunica-ción degrees from the Universitat Politècnica de València, València,Spain, in 1993 and 1997. In 1993he joined the Universitat Politècnica de València, where he is Full Profes-sor since 2003. In 1995 and 1996

he was held a Spanish Trainee position with the European Space research and Technology Centre (ESTEC)-European Space Agency (ESA). He has served on the Editorial Bo-ards of the IEEE Transactions on Microwave Theory and Techniques. His current research interests include nume-rical methods for the analysis of waveguide and scatte-ring structures, automated design of waveguide components, radiating systems, measurement techni-ques, and power effects in passive waveguide systems.

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Waves - 2012 - year 4

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