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Texas Instruments IncorporatedAmplifiers: Op Amps

Expanding the usability of current-feedback amplifiers

By Randy Stephens (Email: r-stephens@)

Systems Specialist, Member Group Technical Staff

Introduction

Although current-feedback (CFB) amplifiers have beenaround as long as the widely utilized voltage-feedback (VFB)amplifiers, their acceptance has been sporadic. One of thereasons for this is quite simple—they have a different

name and therefore must be difficult and very hard to use.This is simply not true. There are numerous papers1, 2, 3comparing the differences between the two amplifiertypes that show they are more similar to each other thandifferent. In fact, for numerous circuits, a CFB amplifiermay actually yield better results due to its inherent slew-rate advantage, lack of a gain-bandwidth product, and reasonably low noise for the performance.

Almost every paper written about CFB amplifiers cautionsreaders that placing a capacitor directly in the feedback path,without any resistance in series, will cause the CFB ampli-fier to oscillate. This is true, as the compensation of theamplifier is tied directly to the feedback impedance. Since acapacitor has low impedance at high frequencies, this essen-tiallyplaces a short in the feedback path that inadvertentlydefeats amplifier compensation, resulting in instability.Because of this limitation, there are a handful of commoncircuits that are not recommended for use with a CFBamplifier. These include integrators, some types of filters,and special feedback-compensation techniques. But whatif there was a way to make these circuits work? And whatif the solution was as simple as adding a single component?This would make it feasible to implement a CFB amplifierfor just about every application for which a VFB amplifiercould be used, with the benefits of the CFB amplifier.

Compensation

This article does not explain the compensation theory ofVFB and CFB amplifiers, as there are many papers writtenon this topic. The only thing that is important is thatthere must be resistance, or impedance, in the feedbackpath at the open-loop intersection point to make the CFBamplifier stable.

Figure 1 shows a traditional VFB amplifier, a THS4012,configured in a noninverting gain of +5 with a simple low-pass gain filter set at approximately 1 MHz by the straight-forward 1/(2πRFCF) formula.

If a CFB amplifier like the THS3112 is simply droppedinto this circuit, it willoscillate and the circuit willbecome useless. A method of compensating the CFB

amplifier in this circuit is to insert a resistance, or imped-ance (Z), in the feedback path as shown in Figure 2.It can easily be seen that regardless of the impedance of the feedback path represented by RFand CF, theimpedance Z is in the amplifier’s feedback loop dictatingthe compensation of the amplifier. The interesting thingabout this configuration is that the feedback resistance(RF), which normally dictates the compensation of the

amplifier, can now be essentially any resistance desired.The reader should keep in mind that this is still a high-speed amplifier with speeds over 100 MHz; so the feedbackresistance should always be kept less than a few kilohmsto minimize the effects of parasitic capacitances on theoverall circuit. Conversely, minimizing the resistance toomuch will place too much of a load on the amplifier, typically degrading performance.

One of the drawbacks of adding the impedance Z in thismanner is that the summing node at the inverting terminalis now separated from the virtual summing node. This can

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introduce errors into the system due to

the bias current and the dynamic signalcurrent flowing through this impedance;but these effects are reasonably small as

long as the impedance is minimized.

Adding impedance Z can affect inputoffset voltage due to the dc input biascurrent, which is typically 1 to 10 µA,

multiplied by the impedance Z. Thisresulting voltage gets multiplied by thenoise gain of the circuit. Additionally, whena signal appears at the output, the CFBamplifier (as the name implies) relies on

an error current flowing through theinvertingnode through the impedance Z,producing a signal error. However, sincethe transimpedance of most CFB ampli-fiers is well over 100 k and sometimes ashigh as several megohms, this error is also

minimized if the impedance is kept low. The

drift of this circuit now also relies on the

temperature characteristics of impedance

Z and should not be used as a precisionamplifier; but most CFB amplifiers are notused as precision amplifiers anyway due to

stated previously. This shows that there is a reasonably widetheir inherent topology limitations. Overall, these issues

range of acceptable values for Z and does not imply that theare minimal and, for most systems, can be effectively

selection for Z is highly critical. Figure 3 also illustrates aignored in favor of the CFB amplifier’s advantages as

common trait for current-feedback amplifiers—as the feed-previously stated.

backimpedance is decreased, the peaking will increase. If

Testing with different Z valuesthe impedance is too low, there is a good chance that theThe easiest way to see if the circuit is stable is to use acircuit will become unstable and oscillate, as illustrated bynetwork analyzer frequency sweep. Instability can typical-the response when Z = 200 .

ly be seen as sharp rises in the frequency response at the

Output noiseamplifier’s bandwidth limitations. If the peaking is smooth,

One element that may be very important in a system is theor there is no peak, then the amplifier should be stable.

output noise. Adding a resistance in the manner discussedFigure 3 shows the frequency response of the system with

only makes the output noise worse. The inverting currentdifferent values of resistors for the variable Z.

noise of the amplifier goes through the resistance at Z andThe response of the THS4012 is also shown for reference

creates a voltage noise. This noise then becomes multipliedto easily compare the performance of the two systems. It

by the circuit’s gain, which is frequency-dependent.is interesting that no matter what resistance is used for Z,

For a CFB amplifier, the inverting current noise is typi-the responses below 20 MHz look identical to each other.

cally the highest noise component of the amplifier. AlthoughThis is the ultimate goal of this configuration—no differ-the CFB amplifier voltage noise is inherently very low, ences in signal performance. For the stability part of the

——

typically less than 3 nV/√Hz, the inverting currentnoise ofcircuit, the area above 20 MHz must be examined.

——

most CFB amplifiers is generally around 15 to 20 pA/√Hz.Examining the circuits in Figures 1 and 2 shows us that

The noninverting current noise is only noticeable if thethe feedback impedance is dictated by the capacitor CF.

source impedance is high. Using a 50- environment Above 20 MHz, this impedance is very small—essentially

minimizes the noninverting current noise.creating a short from the output to the summing node. This

The THS3112 was designed to have very low noise. Theconfiguration is commonly referred to as a unity buffer with

——4voltage noise is 2.2 nV/√Hz, the noninverting current noisethe signal gain set to 1. The data sheet for the THS3112

——

is 2.9 pA/√Hz, and the critical inverting current noise is arecommends that, in a gain of +1 under the circuit condi-——

low 10.8 pA/√Hz. However, multiplying the inverting currenttions utilized, the feedback resistance be 1 k . Thus, it is

noise by 1 k and then multiplying by the gain can aloneno surprise to see that when Z = 1 k , the response looks

——

produce a very substantial output noise of about 54 nV/√Hzvery smooth and well behaved, indicating a very stable

in the pass band. To quantify the output noise of the system,system. However, when Z = 681 , the response also looks

the circuits shown in Figures 1 and 2 were tested for outputvery reasonable and helps minimize the potential issues

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noise (see Figure 4). For comparison, the THS4012, with a

——

respectable voltage noise of 7.5 nV/√Hzand both current

——

noises of 1 pA/√Hz, is also shown in Figure 4.

Note that the output noise of the THS4012 is the sameas when using the THS3112 with Z = 475 . Again, theseresponses are just like those of a VFB amplifier in the tradi-tionalconfiguration, showing that the basic functionality issound—thereare no differences between a VFB amplifierand this configuration. Figure 4 shows that although usingZ = 1 k produces a very stable amplifier, the output

——

noise is 20 nV/√Hzhigher than that of the THS4012.

Keep in mind that the THS3112 has very low overallnoise but that many other CFB amplifiers will probablyproduce much higher noise. The only way to get aroundthis is if the unity-gain stability of the amplifier requires avery small resistor of, say, only 500 or less. But what ifthere was another way to make the CFB amplifier stableandhave low noise at the same time?

Fundamentally speaking, the circuit needs high impedancewithin the feedback path only at the amplifier’s bandwidthlimit. At frequencies below this point, it really does notmatter what the impedance is, and the amplifier will work

fine. The issues stated previously are alsominimized, resulting in an even bettersystem than one using pure resistors.The first solution that comes to mind isto use an inductor. Inductors have lowimpedance at low frequencies and highimpedance at high frequencies—exactlywhat is desired; but their relatively largesize and high cost are generally consideredprohibitive. An alternative componentthat minimizes these disadvantages andstill functions the same is the ferrite chip.

Testing with ferrite chips used for Z

Ferrite chips have been available for severalyears, are relatively low-cost, and areavailable in verysmall sizes—0402 andlarger. Although several manufacturersproduce ferrite chips, testing was donewith what was availablein the test lab—ferrite chips from Murata’s BLM series.Examining the impedance characteristicsof these ferrites revealed several possiblecomponents that could be utilized.

The first factor in determining the propercomponent was the ferrite’s impedance atthe amplifier’s bandwidth limit. For theTHS3112, this implied an impedance of at least 600 at about 150 MHz to meetstability. This can vary, as the first testresults showed (see Figure 3).

Additionally, the Q of the ferrite chipsvaries from grade to grade. Some have alow Q with a fairly smooth rise to the resonance point that then subsides due toinherent properties and parasitics, whileother chips have a relatively high Q with asharp rise and fall in impedance associatedwith them. Although either style may

meet the impedance requirements, testingwas required to see if this Q had an effecton the circuit. Again, the best way to showthe results was to graph the frequencyresponse of the system, as shown inFigure 5. The responses below 10 MHzwere all identical to the original configu-ration. This figure concentrates on thestability portion of the responses above 10 MHz. For comparison purposes, the681- , pure-resistance response is shown.

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Although all of these ferrite chips have

the same impedance at 100 MHz (600 ),they produced different results. The HDseries high-Q chip shows a very narrow

and large peak that will most likely result

in instability and oscillations. The AG andHG series low-Q chips both performed

about the same, and either one would

probably produce acceptable results. The

only difference is that the HG series has

impedance at higher frequencies andwould probably be better suited for usewith very high-speed CFB amplifiers suchas the OPA685 or the THS3202.

Notice that the pure resistance has a

lower response peak than the ferrite chips.

Coupled with the fact that the HD series

has a high Q and a high peak, this impliesthat the slope of the impedance at the

amplifier’s bandwidth is a factor for stabil-ity. This makes a lot of sense; as it is well

known that for any amplifier, if a zerointersects the amplifier’s open-loopresponse at a rate of closure of 40 dB/decade, large peaking and oscillations willmost likely result.5For this circuit config-uration, if the impedance of Z has a largeslope that intersects the transimpedancecurve at essentially a rate of closure of 40 dB/decade, peaking and oscillationsalso will most likely occur. By comparison,a resistor intersects the transimpedancecurve at a rate of closure of 20 dB/decade,resulting in a stable response. Even though

the low-Q ferrite beads have some slope

related to their impedance, the rate ofclosure is much lower than 40 dB/decade,

providing improved stability. Nevertheless,

minimizing this intersection rate of closureas much as possible should produceacceptable results.To further expand on the usefulness of

the ferrite chips, more testing was doneutilizing the AG series in the circuit,asshown in Figure 6.

This figure shows that, just like the

results for the pure resistor, the higher

the impedance is, the lower the peaking.

How does this affect the output noise ofthe system? Figure 7 shows the outputnoise when the ferrite chips were used,along with the output noise of the THS4012

and some of the original resistor configurations.Inverting gain configurationAs expected, due to the low frequency impedance of theAll of the testing discussed so far was done with the non-ferrite chips, the noise is extremely low. This noise was theinverting gain configuration. This configuration forces thesame regardless of which ferrite was used. If noise aboveinverting node voltage to move proportionally to the input10 MHz was important, the impedance of these ferritevoltage applied. So how does the system work in thechips would start to increase the output noise to the sameinverting gain configuration where the inverting node isextent as resistors. These tests show that there are severalheld at a virtual ground? The easy answer is that it worksadvantages of using ferrite chips over resistors.

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exactly the same as before. Figures 8 and 9 show the testcircuits for this configuration. The signal gain was kept ata gain of 5.

The same concepts apply for this CFB configuration asfor the noninverting configuration. The advantage of thiscircuit is that the attenuation is not limited to unity gain,or 0 dB, like the noninverting gain circuit. Figure 10 showsthe frequency responses of this configuration with varyingpure resistor values for Z. The THS4012 response is shownfor comparison purposes.

As expected, the responses all look comparable to eachother below 10 MHz. Additionally, the resistance valuesaffect the stability and again show that the higher theresistance is, the better the stability. Using a resistance aslow as 475 actually shows respectable performance in thisconfiguration. Remember that for oscillations to occur, thegain must be above unity gain, or 0 dB. As long as the peakis below 0 dB, oscillations should not occur. As in the non-inverting case, using 200 shows a large narrow peak thatwill most likely result in stability issues and/or oscillations.However, notice that above 10 MHz the same generalshape occurs for both the CFB and VFB amplifiers. This iscaused by the amplifiers’ input and output impedancesbecoming very high above their bandwidth limit. Whenthis occurs, there is a path for the input signal to flowthrough RG, through CF, and then to feed forward to theload. Of course, the amplifiers’ own input and output

capacitances also affect the amount of feed-through in thecircuit; but it is important to remember that this occursabove the amplifiers’ usable bandwidths.

Just as for the noninverting configuration, using ferritechips has several advantages for the inverting configuration.

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Figure 11 shows the frequency responsesof several of these chips. Figure 12 showsthe results of using various ferrite chipsfrom the same AG family.

As expected, all of these graphs showthe same type of results obtained withthe noninverting configuration. Using alow-Q ferrite chip with high impedancewill result in a stable system. Although thenoise plots for this configuration are notpresented here, they will show the sametype of results as the noninverting configu-ration; using ferrite chips will have thelowest output noise of any configuration.

Conclusion

Although this article shows only two con-figurations with capacitors in the feedbackpath, it shows the fundamental feasibilityof this compensation technique. Whileresistors do work very well, producing themost stable responses, the drawbacks ofthe output noise coupled with the dc andac errors may limit some of the ing ferrite chips helps alleviate manyof these issues, producing the lowest noiseof all with no dc errors or in-band ac sig-nal errors; and stability is almost as goodas when utilizing resistors. It is importantto choose the proper ferrite chip with theamplifier; but this is considered normalprocedure for any circuit design and is nomore difficult than selecting the rightamplifier for the system.

This simple technique helps eliminateone of the major drawbacks of using theCFB amplifier while allowing any systemto enjoy many of its benefits. Designers ofmultiple feedback filters, for example, oncelimited to the use of VFB amplifiers, cannow take advantage of the superior slewrates and lack of gain-bandwidth productcharacteristics found in the CFB amplifier.

References

For more information related to this article,you can downloadan Acrobat Reader fileat /sc/techlit/litnumberandreplace “litnumber”with the TI Lit. #for the materials listed below.

Document TitleTI Lit. #1.“Voltage Feedback Vs. Current Feedback

Op Amps,” Application Report . . . . . . . . . . . . . .slva0512.“The Current-Feedback Op Amp: A High-Speed Building Block,” Application Bulletin . . .sboa0763.“Current Feedback Amplifiers: Review, Stability Analysis, and Applications,”

Application Bulletin . . . . . . . . . . . . . . . . . . . . . . .sboa0814.“Low-Noise, High-Speed Current Feedback

Amplifiers,” Data Sheet . . . . . . . . . . . . . . . . . . . .slos3855.“Effect of Parasitic Capacitance in Op Amp

Circuits,” Application Report . . . . . . . . . . . . . . .sloa013

Related Web sites

/sc/device/partnumber

Replace partnumberwith OPA685, THS3112, THS3202orTHS4012

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